2.7 V, 800 μA, 80 MHz Rail-to-Rail I/O Amplifiers

Data Sheet

AD8031/AD8032


8

FEATURES CONNECTION DIAGRAMS

Low power

Supply current 800 μA/amplifier

Fully specified at +2.7 V, +5 V, and ±5 V supplies High speed and fast settling on 5 V


NC

–IN

+IN

–VS


1

7

2

6

3

+

5

4

AD8031

NC = NO CONNECT


NC

+VS OUT NC


OUT1 1

–IN1 2

3

01056-001

+IN1

–VS 4


AD8032


+ –


8 +VS

7 OUT2

01056-002

6 –IN2

5 +IN2

80 MHz, −3 dB bandwidth (G = +1) 30 V/μs slew rate

125 ns settling time to 0.1%

Figure 1. 8-Lead PDIP (N) and SOIC_N (R)

Figure 2. 8-Lead PDIP (N), SOIC_N (R), and MSOP (RM)

Rail-to-rail input and output

No phase reversal with input 0.5 V beyond supplies


VOUT 1

AD8031


5 +VS

Input CMVR extends beyond rails by 200 mV Output swing to within 20 mV of either rail

Low distortion

–VS 2


+IN 3


+ –

01056-003

4 –IN

−62 dB @ 1 MHz, VO = 2 V p-p

−86 dB @ 100 kHz, VO = 4.6 V p-p

Output current: 15 mA

High grade option: VOS (maximum) = 1.5 mV


APPLICATIONS

High speed, battery-operated systems High component density systems Portable test instruments

A/D buffers Active filters

High speed, set-and-demand amplifiers

Figure 3. 5-Lead SOT-23 (RJ-5)


Operating on supplies from +2.7 V to +12 V and dual supplies up to ±6 V, the AD8031/AD8032 are ideal for a wide range of applications, from battery-operated systems with large bandwidth requirements to high speed systems where component density requires lower power dissipation. The AD8031/AD8032 are available in 8-lead PDIP and 8-lead SOIC_N packages and operate over the industrial temperature range of −40°C to

VIN = 4.85V p-p

+85°C. The AD8031A is also available in the space-saving 5-lead SOT-23 package, and the AD8032A is available in an 8-lead MSOP package.


GENERAL DESCRIPTION

The AD8031 (single) and AD8032 (dual) single-supply, voltage feedback amplifiers feature high speed performance with

80 MHz of small signal bandwidth, 30 V/μs slew rate, and 125 ns settling time. This performance is possible while consuming less than 4.0 mW of power from a single 5 V supply. These features


2µs/DIV


2µs/DIV


1V/DIV

1V/DIV

VOUT = 4.65V p-p G = +1

increase the operation time of high speed, battery-powered systems without compromising dynamic performance.

Figure 4. Input VIN Figure 5. Output VOUT


01056-004

01056-005

+5V


1kΩ

The products have true single-supply capability with rail-to-rail

input and output characteristics and are specified for +2.7 V, +5 V,


VOUT

and ±5 V supplies. The input voltage range can extend to 500 mV beyond each rail. The output voltage swings to within 20 mV of

VIN +

1.7pF


01056-006

+2.5V

each rail providing the maximum output dynamic range.

The AD8031/AD8032 also offer excellent signal quality for only 800 μA of supply current per amplifier; THD is −62 dBc with a 2 V p-p, 1 MHz output signal, and –86 dBc for a 100 kHz,

4.6 V p-p signal on +5 V supply. The low distortion and fast settling time make them ideal as buffers to single-supply ADCs.

Figure 6. Rail-to-Rail Performance at 100 kHz


Rev. G Document Feedback

Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibilityis assumedby Analog Devicesforitsuse, norforanyinfringements ofpatents orother rightsofthirdpartiesthatmayresultfromitsuse.Specificationssubjecttochangewithoutnotice.No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarksandregisteredtrademarksarethepropertyoftheirrespectiveowners.


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TABLE OF CONTENTS

Features 1

Applications 1

General Description 1

Connection Diagrams 1

Revision History 2

Specifications 3

+2.7 V Supply 3

+5 V Supply 4

±5 V Supply 5

Absolute Maximum Ratings 6

Maximum Power Dissipation 6

ESD Caution 6

Typical Performance Characteristics 7

Theory of Operation 13

Input Stage Operation 13

Overdriving the Input Stage 13

Output Stage, Open-Loop Gain and Distortion vs. Clearance from Power Supply 14

Output Overdrive Recovery 14

Driving Capacitive Loads 15

Applications 16

A 2 MHz Single-Supply, Biquad Band-Pass Filter 16

High Performance, Single-Supply Line Driver 16

Outline Dimensions 18

Ordering Guide 20

REVISION HISTORY

3/14—Rev. F to Rev. G

Changes to Second Paragraph of Theory of Operation Section... 13 Changes to Ordering Guide 20

8/13—Rev. E to Rev. F

Changed Input Current Noise at f = 100 kHz from 2.4 pA/√Hz to 0.4 pA/√Hz (Throughout) 3

6/13—Rev. D to Rev. E

Changes to DC Performance Parameter, Table 1 3

Updated Outline Dimensions 19

Changes to Ordering Guide 20

11/08—Rev. C to Rev. D

Change to Table 3 Column Heading 5

Change to Ordering Guide 20

7/06—Rev. B to Rev. C

Updated Format..................................................................Universal

Updated Outline Dimensions 18

Change to Ordering Guide 20

9/99—Rev. A to Rev. B


SPECIFICATIONS

+2.7 V SUPPLY

@ TA = 25°C, VS = 2.7 V, RL = 1 kΩ to 1.35 V, RF = 2.5 kΩ, unless otherwise noted.

Table 1.


Parameter


Conditions

AD8031A/AD8032A

Min Typ Max

AD8031B/AD8032B

Min Typ Max


Unit

DYNAMIC PERFORMANCE

–3 dB Small Signal Bandwidth Slew Rate

Settling Time to 0.1%


G = +1, VO < 0.4 V p-p G = −1, VO = 2 V step

G = −1, VO = 2 V step, CL = 10 pF


54

25


80

30

125


54

25


80

30

125


MHz

V/µs ns

DISTORTION/NOISE PERFORMANCE





Total Harmonic Distortion

fC = 1 MHz, VO = 2 V p-p, G = +2

−62

−62

dBc


fC = 100 kHz, VO = 2 V p-p, G = +2

−86

−86

dBc

Input Voltage Noise

f = 1 kHz

15

15

nV/√Hz

Input Current Noise

f = 100 kHz

0.4

0.4

pA/√Hz


f = 1 kHz

5

5

pA/√Hz

Crosstalk (AD8032 Only)

f = 5 MHz

−60

−60

dB

DC PERFORMANCE









Input Offset Voltage

VCM = VCC/2; VOUT = 1.35 V


±1

±6


±0.5

±1.5

mV


TMIN to TMAX


±6

±10


±1.6

±2.5

mV

Offset Drift

VCM = VCC/2; VOUT = 1.35 V


10



10


µV/°C

Input Bias Current

VCM = VCC/2; VOUT = 1.35 V


0.45

2


0.45

2

µA


TMIN to TMAX



2.2



2.2

µA

Input Offset Current



50

500


50

500

nA

Open-Loop Gain

VCM = VCC/2; VOUT = 0.35 V to 2.35 V

76

80


76

80


dB


TMIN to TMAX

74



74



dB

INPUT CHARACTERISTICS









Common-Mode Input Resistance



40



40


Differential Input Resistance



280



280


Input Capacitance



1.6



1.6


pF

Input Voltage Range



−0.5 to



−0.5 to


V




+3.2



+3.2



Input Common-Mode Voltage Range



−0.2 to



−0.2 to


V




+2.9



+2.9



Common-Mode Rejection Ratio

VCM = 0 V to 2.7 V

46

64


46

64


dB


VCM = 0 V to 1.55 V

58

74


58

74


dB

Differential Input Voltage




3.4



3.4

V

OUTPUT CHARACTERISTICS









Output Voltage Swing Low

RL = 10 kΩ

0.05

0.02


0.05

0.02


V

Output Voltage Swing High


2.6

2.68


2.6

2.68


V

Output Voltage Swing Low

RL = 1 kΩ

0.15

0.08


0.15

0.08


V

Output Voltage Swing High


2.55

2.6


2.55

2.6


V

Output Current



15



15


mA

Short Circuit Current

Sourcing


21



21


mA


Sinking


−34



−34


mA

Capacitive Load Drive

G = +2 (See Figure 46)


15



15


pF

POWER SUPPLY









Operating Range


2.7


12

2.7


12

V

Quiescent Current per Amplifier



750

1250


750

1250

μA

Power Supply Rejection Ratio

VS− = 0 V to −1 V or VS+ = +2.7 V to +3.7 V

75

86


75

86


dB


+5 V SUPPLY

@ TA = 25°C, VS = 5 V, RL = 1 kΩ to 2.5 V, RF = 2.5 kΩ, unless otherwise noted.

Table 2.


Parameter


Conditions

AD8031A/AD8032A

Min Typ Max

AD8031B/AD8032B

Min Typ Max


Unit

DYNAMIC PERFORMANCE

−3 dB Small Signal Bandwidth Slew Rate

Settling Time to 0.1%


G = +1, VO < 0.4 V p-p G = −1, VO = 2 V step

G = −1, VO = 2 V step, CL = 10 pF


54

27


80

32

125


54

27


80

32

125


MHz

V/µs ns

DISTORTION/NOISE PERFORMANCE





Total Harmonic Distortion

fC = 1 MHz, VO = 2 V p-p, G = +2

−62

−62

dBc


fC = 100 kHz, VO = 2 V p-p, G = +2

−86

−86

dBc

Input Voltage Noise

f = 1 kHz

15

15

nV/√Hz

Input Current Noise

f = 100 kHz

0.4

0.4

pA/√Hz


f = 1 kHz

5

5

pA/√Hz

Differential Gain

RL = 1 kΩ

0.17

0.17

%

Differential Phase

RL = 1 kΩ

0.11

0.11

Degrees

Crosstalk (AD8032 Only)

f = 5 MHz

−60

−60

dB

DC PERFORMANCE









Input Offset Voltage

VCM = VCC/2; VOUT = 2.5 V


±1

±6


±0.5

±1.5

mV


TMIN to TMAX


±6

±10


±1.6

±2.5

mV

Offset Drift

VCM = VCC/2; VOUT = 2.5 V


5



5


µV/°C

Input Bias Current

VCM = VCC/2; VOUT = 2.5 V


0.45

1.2


0.45

1.2

µA


TMIN to TMAX



2.0



2.0

µA

Input Offset Current



50

350


50

250

nA

Open-Loop Gain

VCM = VCC/2; VOUT = 1.5 V to 3.5 V

76

82


76

82


dB


TMIN to TMAX

74



74



dB

INPUT CHARACTERISTICS









Common-Mode Input Resistance



40



40


Differential Input Resistance



280



280


Input Capacitance



1.6



1.6


pF

Input Voltage Range



−0.5 to



−0.5 to


V




+5.5



+5.5



Input Common-Mode Voltage Range



−0.2 to



−0.2 to


V




+5.2



+5.2



Common-Mode Rejection Ratio

VCM = 0 V to 5 V

56

70


56

70


dB


VCM = 0 V to 3.8 V

66

80


66

80


dB

Differential Input Voltage




3.4



3.4

V

OUTPUT CHARACTERISTICS









Output Voltage Swing Low

RL = 10 kΩ

0.05

0.02


0.05

0.02


V

Output Voltage Swing High


4.95

4.98


4.95

4.98


V

Output Voltage Swing Low

RL = 1 kΩ

0.2

0.1


0.2

0.1


V

Output Voltage Swing High


4.8

4.9


4.8

4.9


V

Output Current



15



15


mA

Short Circuit Current

Sourcing


28



28


mA


Sinking


−46



−46


mA

Capacitive Load Drive

G = +2 (See Figure 46)


15



15


pF

POWER SUPPLY









Operating Range


2.7


12

2.7


12

V

Quiescent Current per Amplifier



800

1400


800

1400

µA

Power Supply Rejection Ratio

VS− = 0 V to −1 V or VS+ = +5 V to +6 V

75

86


75

86


dB


±5 V SUPPLY

@ TA = 25°C, VS = ±5 V, RL = 1 kΩ to 0 V, RF = 2.5 kΩ, unless otherwise noted.

Table 3.


Parameter


Conditions

AD8031A/AD8032A

Min Typ Max

AD8031B/AD8032B

Min Typ Max


Unit

DYNAMIC PERFORMANCE

−3 dB Small Signal Bandwidth Slew Rate

Settling Time to 0.1%


G = +1, VO < 0.4 V p-p G = −1, VO = 2 V step

G = −1, VO = 2 V step, CL = 10 pF


54

30


80

35

125


54

30


80

35

125


MHz

V/µs ns

DISTORTION/NOISE PERFORMANCE





Total Harmonic Distortion

fC = 1 MHz, VO = 2 V p-p, G = +2

−62

−62

dBc


fC = 100 kHz, VO = 2 V p-p, G = +2

−86

−86

dBc

Input Voltage Noise

f = 1 kHz

15

15

nV/√Hz

Input Current Noise

f = 100 kHz

0.4

0.4

pA/√Hz


f = 1 kHz

5

5

pA/√Hz

Differential Gain

RL = 1 kΩ

0.15

0.15

%

Differential Phase

RL = 1 kΩ

0.15

0.15

Degrees

Crosstalk (AD8032 Only)

f = 5 MHz

−60

−60

dB

DC PERFORMANCE









Input Offset Voltage

VCM = 0 V; VOUT = 0 V


±1

±6


±0.5

±1.5

mV


TMIN to TMAX


±6

±10


±1.6

±2.5

mV

Offset Drift

VCM = 0 V; VOUT = 0 V


5



5


µV/°C

Input Bias Current

VCM = 0 V; VOUT = 0 V


0.45

1.2


0.45

1.2

µA


TMIN to TMAX



2.0



2.0

µA

Input Offset Current



50

350


50

250

nA

Open-Loop Gain

VCM = 0 V; VOUT = ±2 V

76

80


76

80


dB


TMIN to TMAX

74



74



dB

INPUT CHARACTERISTICS









Common-Mode Input Resistance



40



40


Differential Input Resistance



280



280


Input Capacitance



1.6



1.6


pF

Input Voltage Range



−5.5 to



−5.5 to


V




+5.5



+5.5



Input Common-Mode Voltage Range



−5.2 to



−5.2 to


V




+5.2



+5.2



Common-Mode Rejection Ratio

VCM = −5 V to +5 V

60

80


60

80


dB


VCM = −5 V to +3.5 V

66

90


66

90


dB

Differential/Input Voltage




3.4



3.4

V

OUTPUT CHARACTERISTICS









Output Voltage Swing Low

RL = 10 kΩ

−4.94

−4.98


−4.94

−4.98


V

Output Voltage Swing High


+4.94

+4.98


+4.94

+4.98


V

Output Voltage Swing Low

RL = 1 kΩ

−4.7

−4.85


−4.7

−4.85


V

Output Voltage Swing High


+4.7

+4.75


+4.7

+4.75


V

Output Current



15



15


mA

Short Circuit Current

Sourcing


35



35


mA


Sinking


−50



−50


mA

Capacitive Load Drive

G = +2 (See Figure 46)


15



15


pF

POWER SUPPLY









Operating Range


±1.35


±6

±1.35


±6

V

Quiescent Current per Amplifier



900

1600


900

1600

µA

Power Supply Rejection Ratio

VS− = −5 V to −6 V or VS+ = +5 V to +6 V

76

86


76

86


dB


ABSOLUTE MAXIMUM RATINGS

Parameter

Rating

Supply Voltage

12.6 V

Internal Power Dissipation1


8-Lead PDIP (N)

1.3 W

8-Lead SOIC_N (R)

0.8 W

8-Lead MSOP (RM)

0.6 W

5-Lead SOT-23 (RJ)

0.5 W

Input Voltage (Common Mode)

±VS ± 0.5 V

Differential Input Voltage

±3.4 V

Output Short-Circuit Duration

Observe Power Derating Curves

Storage Temperature Range (N, R, RM, RJ)

−65°C to +125°C

Lead Temperature (Soldering 10 sec)

300°C

Table 4.


1 Specification is for the device in free air: 8-Lead PDIP: θJA = 90°C/W.

8-Lead SOIC_N: θJA = 155°C/W.

8-Lead MSOP: θJA = 200°C/W.

5-Lead SOT-23: θJA = 240°C/W.


MAXIMUM POWER DISSIPATION

The maximum power that can be safely dissipated by the AD8031/AD8032 is limited by the associated rise in junction temperature. The maximum safe junction temperature for plastic encapsulated devices is determined by the glass transition temperature of the plastic, approximately 150°C. Exceeding this limit temporarily can cause a shift in parametric performance due to a change in the stresses exerted on the die by the package. Exceeding a junction temperature of 175°C for an extended period can result in device failure.

While the AD8031/AD8032 are internally short-circuit protected, this may not be sufficient to guarantee that the maximum junction temperature (150°C) is not exceeded under all conditions. To ensure proper operation, it is necessary to observe the maximum power derating curves shown in Figure 7.

2.0

TJ = +150°C

MAXIMUM POWER DISSIPATION (W)

8-LEAD PDIP

Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect


1.5


1.0


8-LEAD MSOP


8-LEAD SOIC

device reliability.

0.5 5-LEAD SOT-23


01056-007

0

–50 –40 –30 –20 –10 0 10 20 30 40 50 60 70 80 90

AMBIENT TEMPERATURE (°C)

Figure 7. Maximum Power Dissipation vs. Temperature


ESD CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality.


TYPICAL PERFORMANCE CHARACTERISTICS

90


800


80

N = 250

NUMBER OF PARTS IN BIN

70


60


50


40


30


20


10


600


INPUT BIAS CURRENT (nA)

400


200


0


–200


–400


–600


VS = 2.7V


VS = 5V


VS = 10V


0

–5 –4 –3 –2 –1 0 1 2 3 4 5

VOS (mV)


–800


01056-011

0 1 2 3 4 5 6 7 8 9 10

COMMON-MODE VOLTAGE (V)

01056-008

Figure 8. Typical VOS Distribution @ VS = 5 V Figure 11. Input Bias Current vs. Common-Mode Voltage


2.5


2.3


0












VS = 5V



















































–0.1



OFFSET VOLTAGE (mV)

2.1


1.9


1.7


VS = +5V


VS = ±5V

–0.2


OFFSET VOLTAGE (mV)

–0.3


–0.4


01056-009

–0.5


1.5

–40 –30 –20 –10 0 10 20 30 40 50 60 70 80 90

TEMPERATURE (°C)


Figure 9. Input Offset Voltage vs. Temperature

–0.6


01056-012

0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0

COMMON-MODE VOLTAGE (V)

Figure 12. VOS vs. Common-Mode Voltage


1.00


0.95


0.90


INPUT BIAS (µA)

0.85


0.80


0.75


0.70


0.65


0.60


0.55























VS = 5V











































































































0.50


1000


SUPPLY CURRENT/AMPLIFIER (µA)

950


900


850


800


750


700


650


600


±IS, VS = ±5V


+IS, VS = +5V


01056-010

+IS, VS = +2.7V

–40 –30 –20 –10 0 10 20 30 40 50 60 70 80 90

TEMPERATURE (°C)

Figure 10. Input Bias Current vs. Temperature

–40 –30 –20 –10 0 10 20 30 40 50 60 70 80 90

01056-013

TEMPERATURE (°C)

Figure 13. Supply Current vs. Temperature


0


DIFFERENCE FROM VCC (V)

–0.5


–1.0


–1.5


VCC = 2.7V


VCC = 5V


VCC

1.2


DIFFERENCE FROM VEE (V)

1.0


0.8


0.6


VCC = 10V


VCC = 5V


VIN


VCC


VEE


R VCC

2


VOUT LOAD


–2.0


–2.5

VCC = 10V


VIN


VEE

R VCC

2

VOUT LOAD

0.4


0.2


0


01056-017

VCC = 2.7V

100 1k 10k RLOAD (Ω)

100

1k RLOAD (Ω)

10k

01056-014

Figure 14. +Output Saturation Voltage vs. RLOAD @ +85°C Figure 17. −Output Saturation Voltage vs. RLOAD @ +85°C


0


DIFFERENCE FROM VCC (V)

–0.5


–1.0


VCC


= 2.7V


VCC = 5V


1.2


DIFFERENCE FROM VEE (V)

1.0


0.8


VCC = 10V


VIN


VCC


VEE


VOUT RLOAD

V


–1.5


–2.0


–2.5


VCC = 10V


VIN


VCC


VEE


R VCC

01056-015

2


VOUT LOAD


0.6


0.4


0.2


0


VCC = 2.7V

CC

2


VCC = 5V

100 1k 10k RLOAD (Ω)

100

1k RLOAD (Ω)

10k

01056-018

Figure 15. +Output Saturation Voltage vs. RLOAD @ +25°C Figure 18. −Output Saturation Voltage vs. RLOAD @ +25°C


0


DIFFERENCE FROM VCC (V)

–0.5


–1.0


VCC = 2.7V


VCC = 5V

1.2


DIFFERENCE FROM VEE (V)

1.0


0.8


VCC = 10V


VIN


VCC


VEE


VOUT RLOAD

V


–1.5


–2.0


–2.5


VCC


= 10V


VIN


VCC


VEE


VOUT RLOAD

01056-016

VCC 2


0.6


0.4


0.2


0

CC

2


VCC = 5V


01056-019

VCC = 2.7V

100 1k 10k RLOAD (Ω)

100

1k RLOAD (Ω)

10k

Figure 16. +Output Saturation Voltage vs. RLOAD @ −40°C Figure 19. −Output Saturation Voltage vs. RLOAD @ −40°C


VS = 5V

110


1V

500mV

105


100

90

–AOL

INPUT BIAS CURRENT (mA)

100

10

95


+AOL

GAIN (dB)

90


85 0


80


75


70


65


60

0 2k 4k 6k 8k 10k

RLOAD (Ω)


–10


VS = 5V

10

0%


500mV


01056-023

–1.5 0.5 2.5 4.5 6.5

INPUT VOLTAGE (V)

01056-020

Figure 20. Open-Loop Gain (AOL) vs. RLOAD Figure 23. Differential Input Overvoltage I-V Characteristics


86


84

–AOL


GAIN (dB)

82


+AOL

80


78


VS = 5V RL = 1kΩ

0.05


DIFF GAIN (%)

0


–0.05


–0.10


–0.15


DIFF PHASE (Degrees)

0.10


0.05


0


–0.05









































































































1ST 2ND 3RD 4TH 5TH 6TH 7TH 8TH 9TH 10TH 11TH


76

–40 –30 –20 –10 0 10 20 30 40 50 60 70 80 90

TEMPERATURE (°C)


–0.10


01056-024

1ST 2ND 3RD 4TH 5TH 6TH 7TH 8TH 9TH 10TH 11TH

01056-021

Figure 21. Open Loop Gain vs. (AOL) Temperature Figure 24. Differential Gain and Phase @ VS = ±5 V; RL = 1 kΩ


110


100


AOL (dB)

90


80

100


INPUT VOLTAGE NOISE (nV/ Hz)

30


10


VOLTAGE NOISE


VS = 5V


INPUT CURRENT NOISE (pA/ Hz)

100


10












VS = 5V

RLOAD = 10kΩ























RLOAD = 1kΩ































3

70

CURRENT NOISE

1

01056-022

60


1


0.1


50

0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0

VOUT (V)

Figure 22. Open-Loop Gain (AOL) vs. VOUT

0.3

01056-025

10 100 1k 10k 100k 1M 10M

FREQUENCY (Hz)


Figure 25. Input Voltage Noise vs. Frequency


5


4 VS = 5V

G = +1

NORMALIZED GAIN (dB)

3 RL = 1kΩ

2


1


0


–1


–2


–3


–4


–5 0.1 1 10 100

FREQUENCY (MHz)


–90


PHASE (Degrees)

–135


–180


–225


0.3


1 10

FREQUENCY (MHz)


100


40


OPEN-LOOPGAIN (dB)






























































GAIN



































































































PHASE
































































































30


20


10


0


–10


–20

01056-026

01056-029

Figure 26. Unity Gain, −3 dB Bandwidth Figure 29. Open-Loop Frequency Response


3


2 VS = 5V

VIN = –16dBm

NORMALIZED GAIN (dB)

1


+85°C

–20


TOTAL HARMONIC DISTORTION (dBc)

–30


G = +1, R = 2kΩ TOVCC


0


–1


–2


–3

VIN

–4


VS

2kΩ

VOUT

50Ω


–40°C

+25°C


–40


–50


–60


–70

L 2

2.5V p-p VS = 2.7V

1.3V p-p VS = 2.7V


2V p-p VS = 2.7V


–5 0.1 1 10 100

FREQUENCY (MHz)


–80

4.8V p-p VS = 5V

1k 10k 100k 1M

FUNDAMENTAL FREQUENCY (Hz)


10M

01056-027

01056-030

Figure 27. Closed-Loop Gain vs. Temperature Figure 30. Total Harmonic Distortion vs. Frequency; G = +1


2

VS = +2.7V

1 RL + CL TO 1.35V

CLOSED-LOOP GAIN (dB)

0


–1


–2


–3

G = +1

–4 CL = 5pF

RL = 1kΩ

–5


–6


–7


VS = ±5V


VS = +5V RL + CL TO 2.5V

–20


TOTAL HARMONIC DISTORTION (dBc)
































G = +2

VS = 5V V CC RL = 1kΩ TO 2






















































4.8V p-p


















































1V p-p




































4.6V p-p































4V p-p































–30


–40


–50


–60


–70


–80


–90


–8

100k


1M 10M 100M FREQUENCY (Hz)


–100

01056-031

1k 10k 100k 1M 10M

FUNDAMENTAL FREQUENCY (Hz)

01056-028

Figure 28. Closed-Loop Gain vs. Supply Voltage

Figure 31. Total Harmonic Distortion vs. Frequency; G +2


10

VS = ±5V


8


OUTPUT (V p-p)

6

VS = +5V


4


VS = +2.7V

01056-032

2

0


POWER SUPPLY REJECTION RATIO (dB)

–20


–40


–60


–80


–100


VS = 5V



0

1k 10k 100k 1M 10M

FREQUENCY (Hz)

Figure 32. Large Signal Response

–120


01056-035

100 1k 10k 100k 1M 10M 100M FREQUENCY (Hz)

Figure 35. PSRR vs. Frequency


100

50


ROUT (Ω)

10


1

RBT = 50Ω



VS = 5V

RL = 10kΩ TO 2.5V VIN = 6V p-p

G = +1

5.5


4.5


1V/DIV

3.5


2.5


1.5


0.1


RBT = 0Ω


RBT

V


OUT


0.5


–0.5


01056-033

0.1 1 10 100 200

FREQUENCY (MHz)

Figure 33. ROUT vs. Frequency


10µs/DIV

01056-036

Figure 36. Output Voltage


COMMON-MODE REJECTION RATIO (dB)

0


–20


–40


–60


01056-034

–80


5.5


4.5


1V/DIV

3.5


2.5


1.5


0.5


–0.5


INPUT


VS = 5V G = +1

INPUT = 650mV BEYOND RAILS




























VS = 5V













































































































–100

100 1k 10k 100k 1M 10M

FREQUENCY (Hz)

Figure 34. CMRR vs. Frequency


10µs/DIV


01056-037

Figure 37. Output Voltage Phase Reversal Behavior


RL TO

+2.5V



G = +1 RF = 0Ω

RL = 2kΩ TO 2.5V

CL = 5pF TO 2.5V VS = 5V

2.56


2.54


20mV/DIV

2.52


500mV/DIV

2.50


2.48


RL TO GND

0


10µs/DIV


VS = +5V RL = 1kΩ

G = –1


2.46


2.44


01056-041

50ns/DIV

01056-038

Figure 38. Output Swing Figure 41. 100 mV Step Response




G = +2

RF = RG = 2.5kΩ RL = 2kΩ

CL = 5pF

VS = 5V

3.1


2.9


200mV/DIV

2.7


2.5


2.3


2.1


1.9

–50


CROSSTALK(dB)

–60


–70


–80


–90


–100


2.5kΩ


VIN


2.5kΩ


01056-039

50Ω


1kΩ


VS = ±2.5V VIN = +10dBm


2.5kΩ 2.5kΩ

VOUT 50Ω


50ns/DIV

Figure 39. 1 V Step Response


0.1

TRANSMITTER RECEIVER

01056-042

1 10 100 200

FREQUENCY (MHz)

Figure 42. Crosstalk vs. Frequency



2.85


2.35


500mV/DIV

1.85


VS = 2.7V RL = 1kΩ G = –1


1.35


0.85


0.35


RL TO 1.35V


RL TO GND


01056-040

10µs/DIV

Figure 40. Output Swing


THEORY OF OPERATION

The AD8031/AD8032 are single and dual versions of high speed, low power, voltage feedback amplifiers featuring an innovative architecture that maximizes the dynamic range capability on the inputs and outputs. The linear input common- mode range exceeds either supply voltage by 200 mV, and the amplifiers show no phase reversal up to 500 mV beyond supply. The output swings to within 20 mV of either supply when driving a light load; 300 mV when driving up to 5 mA.

Fabricated on Analog Devices, Inc. eXtra Fast Complementary Bipolar (XFCB) process, the amplifier provides an impressive 80 MHz bandwidth when used as a follower and a 30 V/µs slew rate at only 800 µA supply current. Careful design allows the amplifier to operate with a supply voltage as low as 2.7 V.

INPUT STAGE OPERATION

A simplified schematic of the input stage appears in Figure 43. For common-mode voltages up to 1.1 V within the positive supply (0 V to 3.9 V on a single 5 V supply), tail current I2 flows through the PNP differential pair, Q13 and Q17. Q5 is cut off; no bias current is routed to the parallel NPN differential pair, Q2 and Q3. As the common-mode voltage is driven within

1.1 V of the positive supply, Q5 turns on and routes the tail current away from the PNP pair and to the NPN pair. During this transition region, the input current of the amplifier changes magnitude and direction. Reusing the same tail current ensures that the input stage has the same transconductance, which determines the gain and bandwidth of the amplifier, in both regions of operation.


Switching to the NPN pair as the common-mode voltage is driven beyond 1 V within the positive supply allows the amplifier to provide useful operation for signals at either end of the supply voltage range and eliminates the possibility of phase reversal for input signals up to 500 mV beyond either power supply. Offset voltage also changes to reflect the offset of the input pair in control. The transition region is small, approximately 180 mV. These sudden changes in the dc parameters of the input stage can produce glitches that adversely affect distortion.

OVERDRIVING THE INPUT STAGE

Sustained input differential voltages greater than 3.4 V should be avoided as the input transistors can be damaged. Input clamp diodes are recommended if the possibility of this condition exists.

The voltages at the collectors of the input pairs are set to

200 mV from the power supply rails. This allows the amplifier to remain in linear operation for input voltages up to 500 mV beyond the supply voltages. Driving the input common-mode voltage beyond that point will forward bias the collector junction of the input transistor, resulting in phase reversal. Sustaining this condition for any length of time should be avoided because it is easy to exceed the maximum allowed input differential voltage when the amplifier is in phase reversal.


Q9

1.1V

R5


VIN


I2 90µA

VCC


Q3 Q2


R1

2kΩ


I3 25µA


Q6 Q10


R2

2kΩ

50kΩ R6 R7 1 1

Q5 850Ω

850Ω R8

850Ω

R9 4 Q8 Q7 4 850Ω

VIP

Q13 Q17


Q14

4

1


Q15

I4 25µA


Q16


Q11

4


1

OUTPUT STAGE, COMMON-MODE FEEDBACK



I1

5µA


VEE


Q18 Q4


R3

2kΩ

R4

01056-043

2kΩ


Figure 43. Simplified Schematic of AD8031 Input Stage


OUTPUT STAGE, OPEN-LOOP GAIN AND DISTORTION vs. CLEARANCE FROM POWER SUPPLY

The AD8031 features a rail-to-rail output stage. The output transistors operate as common-emitter amplifiers, providing the output drive current as well as a large portion of the amplifier’s open-loop gain.

The open-loop gain of the AD8031 decreases approximately linearly with load resistance and depends on the output voltage. Open-loop gain stays constant to within 250 mV of the positive power supply, 150 mV of the negative power supply, and then decreases as the output transistors are driven further into saturation.

The distortion performance of the AD8031/AD8032 amplifiers differs from conventional amplifiers. Typically, the distortion


DIFFERENTIAL

DRIVE FROM

INPUT STAGE

I1

25µA

Q42 Q51


Q37 Q38 R29


Q68

I2

25µA


C9

+

5pF


Q47

performance of the amplifier degrades as the output voltage amplitude increases.

Used as a unity gain follower, the output of the AD8031/ AD8032 exhibits more distortion in the peak output voltage

Q20


Q21

300Ω


Q27

region around VCC − 0.7 V. This unusual distortion characteristic is


I4 25µA


+

Q43 Q48


I5 25µA

C5 1.5pF


Q49

VOUT

caused by the input stage architecture and is discussed in detail in the Input Stage Operation section,

OUTPUT OVERDRIVE RECOVERY

Q50

Q44

Output overdrive of an amplifier occurs when the amplifier

01056-044

attempts to drive the output voltage to a level outside its normal

Figure 44. Output Stage Simplified Schematic


RL

50Ω

The output voltage limit depends on how much current the output transistors are required to source or sink. For applications with low drive requirements (for instance, a unity gain follower driving another amplifier input), the AD8031 typically swings within 20 mV of either voltage supply. As the required current load increases, the saturation output voltage increases linearly as

range. After the overdrive condition is removed, the amplifier must recover to normal operation in a reasonable amount of time. As shown in Figure 45, the AD8031/AD8032 recover within 100 ns from negative overdrive and within 80 ns from positive overdrive.


RF = RG = 2kΩ

RG

RF


VOUT

ILOAD

where:

× RC

VIN

ILOAD is the required load current.

RC is the output transistor collector resistance.

For the AD8031, the collector resistances for both output transistors are typically 25 Ω. As the current load exceeds the rated output current of 15 mA, the amount of base drive current required to drive the output transistor into saturation reaches its limit, and the amplifier’s output swing rapidly decreases.



1V

VS = ±2.5V VIN = ±2.5V

RL = 1kΩ TO GND

100ns


01056-045

Figure 45. Overdrive Recovery


DRIVING CAPACITIVE LOADS


1000


VS = 5V


RS = 5Ω

Capacitive loads interact with an op amp’s output impedance to create an extra delay in the feedback path. This reduces circuit stability and can cause unwanted ringing and oscillation. A given value of capacitance causes much less ringing when the amplifier is used with a higher noise gain.

The capacitive load drive of the AD8031/AD8032 can be increased by adding a low valued resistor in series with the capacitive load. Introducing a series resistor tends to isolate the capacitive load from the feedback loop, thereby diminishing its


100


CAPACITIVE LOAD (pF)

10

200mV STEP

WITH 30% OVERSHOOT


RS = 20Ω


RS = 0Ω, 5Ω


RS = 20Ω


RG RF

RS


RS = 0Ω


VOUT CL

influence. Figure 46 shows the effects of a series resistor on the capacitive drive for varying voltage gains. As the closed-loop gain is increased, the larger phase margin allows for larger capacitive loads with less overshoot. Adding a series resistor at lower closed-loop gains accomplishes the same effect. For large capacitive loads, the frequency response of the amplifier is dominated by the roll-off of the series resistor and capacitive load.

1

01056-046

0 1 2 3 4 5

CLOSED-LOOP GAIN (V/V)

Figure 46. Capacitive Load Drive vs. Closed-Loop Gain


APPLICATIONS

A 2 MHz SINGLE-SUPPLY, BIQUAD BAND-PASS FILTER

Figure 47 shows a circuit for a single-supply, biquad band-pass filter with a center frequency of 2 MHz. A 2.5 V bias level is easily created by connecting the noninverting inputs of all three op amps to a resistor divider consisting of two 1 kΩ resistors connected between 5 V and ground. This bias point is also decoupled to ground with a 0.1 µF capacitor. The frequency response of the filter is shown in Figure 48.

To maintain an accurate center frequency, it is essential that the op amp have sufficient loop gain at 2 MHz. This requires the choice of an op amp with a significantly higher unity gain, crossover frequency. The unity gain, crossover frequency of the AD8031/AD8032 is 40 MHz. Multiplying the open-loop gain by the feedback factors of the individual op amp circuits yields the loop gain for each gain stage. From the feedback networks of the individual op amp circuits, it can be seen that each op amp has a loop gain of at least 21 dB. This level is high enough to ensure that the center frequency of the filter is not affected by the op amp’s bandwidth. If, for example, an op amp with a gain bandwidth product of 10 MHz was chosen in this application, the resulting center frequency would shift by 20% to 1.6 MHz.

R6

1kΩ


0


–10


GAIN (dB)

–20


–30


–40


01056-048

–50

10k 100k 1M 10M 100M FREQUENCY (Hz)

Figure 48. Frequency Response of 2 MHz Band-Pass Filter


HIGH PERFORMANCE, SINGLE-SUPPLY LINE DRIVER

Even though the AD8031/AD8032 swing close to both rails, the AD8031 has optimum distortion performance when the signal has a common-mode level half way between the supplies and when there is about 500 mV of headroom to each rail. If low distortion is required in single-supply applications for signals that swing close to ground, an emitter-follower circuit can be used at the op amp output.


R2

2kΩ

C1 50pF


R4

2kΩ


5V

10µF


VIN


R1

3kΩ


1kΩ

5V

0.1µF


AD8031


R3

2kΩ


5V

0.1µF


1/2


R5

2kΩ


C2 50pF


VIN


49.9Ω

0.1µF


3 7

6

2

4 AD8031


2N3904


0.1µF


1kΩ


VOUT

AD8032

1/2

01056-049

AD8032

2.49kΩ

2.49kΩ 49.9Ω


01056-047

200Ω


49.9Ω

VOUT

Figure 47. A 2 MHz, Biquad Band-Pass Filter Using AD8031/AD8032

Figure 49. Low Distortion Line Driver for Single-Supply, Ground Referenced Signals


Figure 49 shows the AD8031 configured as a single-supply, gain- of-2 line driver. With the output driving a back-terminated

50 Ω line, the overall gain from VIN to VOUT is unity. In addition to minimizing reflections, the 50 Ω back termination resistor protects the transistor from damage if the cable is short circuited. The emitter follower, which is inside the feedback loop, ensures that the output voltage from the AD8031 stays about 700 mV above ground. Using this circuit, low distortion is attainable even when the output signal swings to within 50 mV of ground. The circuit was tested at 500 kHz and 2 MHz.

Figure 50 and Figure 51 show the output signal swing and frequency spectrum at 500 kHz. At this frequency, the output signal (at VOUT), which has a peak-to-peak swing of 1.95 V (50 mV to 2 V), has a THD of −68 dB (SFDR = −77 dB).


100

90


2V


10

0%

01056-050

50mV


This circuit could also be used to drive the analog input of a single-supply, high speed ADC whose input voltage range is referenced to ground (for example, 0 V to 2 V or 0 V to 4 V). In this case, a back termination resistor is not necessary (assuming a short physical distance from transistor to ADC); therefore, the emitter of the external transistor would be connected directly to the ADC input. The available output voltage swing of the circuit would therefore be doubled.


1.5V


100

90


0.5V

1µs


Figure 50. Output Signal Swing of Low Distortion Line Driver at 500 kHz

VERTICAL SCALE (10dB/DIV)





































































































+9dBm


10

0%


0.2V

200ns

01056-052

50mV


Figure 52. Output Signal Swing of Low Distortion Line Driver at 2 MHz





































































































+7dBm


01056-051

VERTICAL SCALE (10dB/DIV)

START 0Hz STOP 5MHz

Figure 51. THD of Low Distortion Line Driver at 500 kHz


Figure 52 and Figure 53 show the output signal swing and frequency spectrum at 2 MHz. As expected, there is some degradation in signal quality at the higher frequency. When the output signal has a peak-to-peak swing of 1.45 V (swinging from 50 mV to 1.5 V), the THD is −55 dB (SFDR = −60 dB).


01056-053

START 0Hz STOP 20MHz

Figure 53. THD of Low Distortion Line Driver at 2 MHz


OUTLINE DIMENSIONS


0.400 (10.16)

0.365 (9.27)

0.355 (9.02)



0.210 (5.33)

MAX

0.150 (3.81)


8

5


1

4


0.100 (2.54) BSC


0.280 (7.11)

0.250 (6.35)

0.240 (6.10)


0.015

(0.38)


0.325 (8.26)

0.310 (7.87)

0.300 (7.62)

0.060 (1.52)

MAX


0.015 (0.38)


0.195 (4.95)

0.130 (3.30)

0.115 (2.92)

0.130 (3.30) MIN

0.115 (2.92) SEATING

GAUGE

PLANE 0.014 (0.36)

0.022 (0.56)

0.018 (0.46)

0.014 (0.36)

PLANE


0.005 (0.13) MIN


0.430 (10.92) MAX

0.010 (0.25)

0.008 (0.20)

0.070 (1.78)

0.060 (1.52)

0.045 (1.14)


COMPLIANT TO JEDEC STANDARDS MS-001

070606-A

CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN. CORNER LEADS MAY BE CONFIGURED AS WHOLE OR HALF LEADS.

Figure 54. 8-Lead Plastic Dual In-Line Package [PDIP] Narrow Body (N-8)

Dimensions shown in inches and (millimeters)


5.00 (0.1968)

4.80 (0.1890)



8 5

4.00 (0.1574)


6.20 (0.2441)

3.80 (0.1497) 1 4 5.80 (0.2284)


1.27 (0.0500) 0.50 (0.0196)


45°


0.25 (0.0098)

0.10 (0.0040)

BSC

1.75 (0.0688)

1.35 (0.0532)

0.25 (0.0099)

COPLANARITY

0.51 (0.0201)


1.27 (0.0500)

0.10

SEATING PLANE

0.31 (0.0122)

0.25 (0.0098)

0.17 (0.0067)

0.40 (0.0157)


COMPLIANT TO JEDEC STANDARDS MS-012-AA

012407-A

CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.

Figure 55. 8-Lead Standard Small Outline Package [SOIC_N] Narrow Body (R-8)

Dimensions shown in millimeters and (inches)


3.00

2.90

2.80



1.70

1.60

1.50

5 4 3.00

2.80

2.60

1 2 3



1.30

1.15

0.90


1.90 BSC

0.95 BSC


1.45 MAX

0.95 MIN


0.20 MAX

0.08 MIN


0.55

0.15 MAX

0.05 MIN


0.50 MAX

0.35 MIN


SEATING PLANE

10°

5°


0.60

BSC

0.45

0.35


11-01-2010-A

COMPLIANT TO JEDEC STANDARDS MO-178-AA

Figure 56. 5-Lead Small Outline Transistor Package [SOT-23] (RJ-5)

Dimensions shown in millimeters


3.20

3.00

2.80



3.20

3.00

2.80

8 5 5.15

4.90

4.65

1 4


PIN 1 IDENTIFIER


0.65 BSC


0.95

0.85

0.75


0.15

0.05

COPLANARITY 0.10


0.40

0.25


1.10 MAX



15° MAX


0.23

0.09


0.80

0.55

10-07-2009-B

0.40


COMPLIANT TO JEDEC STANDARDS MO-187-AA

Figure 57. 8-Lead Mini Small Outline Package [MSOP] (RM-8)

Dimensions shown in millimeters


ORDERING GUIDE

Model1

Temperature Range

Package Description

Package Option

Branding

AD8031ANZ

–40°C to +85°C

8-Lead PDIP

N-8


AD8031AR

–40°C to +85°C

8-Lead SOIC_N

R-8


AD8031ARZ

–40°C to +85°C

8-Lead SOIC_N

R-8


AD8031ARZ-REEL

–40°C to +85°C

8-Lead SOIC_N, 13" Tape and Reel

R-8


AD8031ARZ-REEL7

–40°C to +85°C

8-Lead SOIC_N, 7" Tape and Reel

R-8


AD8031ART-R2

–40°C to +85°C

5-Lead SOT-23

RJ-5

H0A

AD8031ART-REEL7

–40°C to +85°C

5-Lead SOT-23, 7" Tape and Reel

RJ-5

H0A

AD8031ARTZ-R2

–40°C to +85°C

5-Lead SOT-23

RJ-5

H04

AD8031ARTZ-REEL

–40°C to +85°C

5-Lead SOT-23, 13" Tape and Reel

RJ-5

H04

AD8031ARTZ-REEL7

–40°C to +85°C

5-Lead SOT-23, 7" Tape and Reel

RJ-5

H04

AD8031BNZ

–40°C to +85°C

8-Lead PDIP

N-8


AD8031BR

–40°C to +85°C

8-Lead SOIC_N

R-8


AD8031BRZ

–40°C to +85°C

8-Lead SOIC_N

R-8


AD8031BRZ-REEL

–40°C to +85°C

8-Lead SOIC_N, 13" Tape and Reel

R-8


AD8031BRZ-REEL7

–40°C to +85°C

8-Lead SOIC_N, 7" Tape and Reel

R-8


AD8031AR-EBZ


8-Lead SOIC Evaluation Board



AD8031ART-EBZ


5-Lead SOT-23 Evaluation Board



AD8032ANZ

–40°C to +85°C

8-Lead PDIP

N-8


AD8032AR AD8032AR-REEL7 AD8032ARZ AD8032ARZ-REEL AD8032ARZ-REEL7 AD8032ARM AD8032ARM-REEL AD8032ARM-REEL7 AD8032ARMZ AD8032ARMZ-REEL AD8032ARMZ-REEL7 AD8032BNZ AD8032BR AD8032BR-REEL7 AD8032BRZ AD8032BRZ-REEL AD8032BRZ-REEL7 AD8032ACHIPS AD8032AR-EBZ

AD8032ARM-EBZ

–40°C to +85°C

–40°C to +85°C

–40°C to +85°C

–40°C to +85°C

–40°C to +85°C

–40°C to +85°C

–40°C to +85°C

–40°C to +85°C

–40°C to +85°C

–40°C to +85°C

–40°C to +85°C

–40°C to +85°C

–40°C to +85°C

–40°C to +85°C

–40°C to +85°C

–40°C to +85°C

–40°C to +85°C

8-Lead SOIC_N

8-Lead SOIC_N, 7" Tape and Reel 8-Lead SOIC_N

8-Lead SOIC_N, 13" Tape and Reel 8-Lead SOIC_N, 7" Tape and Reel 8-Lead MSOP

8-Lead MSOP, 13" Tape and Reel 8-Lead MSOP, 7" Tape and Reel 8-Lead MSOP

8-Lead MSOP, 13" Tape and Reel 8-Lead MSOP, 7" Tape and Reel 8-Lead PDIP

8-Lead SOIC_N

8-Lead SOIC_N, 7" Tape and Reel 8-Lead SOIC_N

8-Lead SOIC_N, 13" Tape and Reel 8-Lead SOIC_N, 7" Tape and Reel Die

8-Lead SOIC Evaluation Board

8-Lead MSOP Evaluation Board

R-8

R-8

R-8

R-8

R-8 RM-8 RM-8 RM-8 RM-8 RM-8 RM-8 N-8

R-8

R-8

R-8

R-8

R-8


H9A H9A H9A H9A# H9A# H9A#

1 Z = RoHS Compliant Part, # denotes lead-free product may be top or bottom marked.


©2014 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners.

D01056-0-3/14(G)

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Analog Devices Inc.:

AD8031ANZ AD8031ARTZ-REEL7 AD8031ARZ AD8031BNZ AD8031BRZ AD8032ANZ AD8032AR AD8032ARMZ AD8032ARMZ-REEL7 AD8032ARZ AD8032BRZ AD8031AR AD8032BR AD8032ARM AD8031BR AD8031ART-REEL7 AD8031ARTZ-REEL AD8031ARZ-REEL AD8031ARZ-REEL7 AD8031BRZ-REEL7 AD8032ARM-REEL7 AD8032ARMZ-REEL AD8032ARZ-REEL AD8032ARZ-REEL7 AD8032BRZ-REEL AD8032BRZ-REEL7 AD8031ARTZ-R2