New multi‐standard dual‐wideband and quad‐wideband asymmetric step impedance resonator filters with wide stop band restriction

New multi‐standard wide band filters with compact sizes are designed for wireless communication devices. The proposed structures realize dual‐wideband and quad‐wideband characteristics by using a new skew‐symmetrical coupled pair of asymmetric stepped impedance resonators, combined with other structures. The first and second dual‐wideband filters realize fractional bandwidths (FBW) of 43.2%/31.9% at the central frequencies (CF) of 1.875/1.63 GHz, and second bandwidths of 580 MHz/1.75 GHz at CF of 5.52/4.46 GHz, respectively. The proposed quad‐band filter realizes its first/second/third/fourth pass bands at CF 2.13/5.25/7.685/9.31 GHz with FBW of 46.0%/11.4%/4.6% and 5.4%, respectively. The wide pass bands are attributed to the mutual coupling of the modified ASIR resonators and their bandwidths are controllable by tuning relative parameters while the wide stop band performance is optimized by the novel interdigital cross coupled line structure and parallel uncoupled microstrip line structure. Moreover, the quad band is generated by introducing the novel defected rectangle structure. These multi‐standard filters are simulated, fabricated and measured, and measured results agree well with both simulated results and theory predictions. The good in‐band and out‐of‐band performances, the miniaturized sizes and simple structures of the proposed filters make them very promising for applications in future multi‐standard wireless communication.

Compared to the traditional stepped impedance resonator (SIR) with two step discontinuities, the asymmetric SIR has only one discontinuity but retains the characteristic of controllable spurious modes.
It has advantages of compact size, less loss, and versatility in design, particularly for high-order BPFs such as dual band, triple band and quad band BPFs because of its inherently higher order resonant modes. It has recently been used to construct high-performance BPFs, [3][4][5][6][7] and in References 6,7, two-stage wide-stopband BPFs using asymmetric SIRs have been proposed. The schemes of distributing spurious resonant frequencies of asymmetric SIR with wide stop band performance are realized. Additionally, with fewer discontinuities, the asymmetric SIR can be easily folded and coupled, and this results in a lower insertion loss and greater size reduction.
On the other hand, with the current increasingly stringent limitations on frequency spectrum resources and the development of advanced multi-standard wireless communication systems, multi-standard internal filters have become a necessity for state-of-the-art multifunction "smart phones" and wireless transceivers for mobile devices. Such filters are generally required to be capable of covering the frequency bands of the Global Positioning System (GPS: band centered at 1.57 GHz), the Global System for Mobile Communication (GSM: 1800/1900 MHz etc.) and the Universal Mobile Telecommunications System (UMTS: 1710-1880/1850-1990/ 1920-2170 MHz etc.). Moreover the ever expanding implementation of the Wireless Local Area Network (WLAN) adds the requirement of a band centered at 2.4 GHz and/or 5.2 GHz. Many dual band filters have recently been designed to satisfy such demanding requirements. However, many of these that are also miniaturized fail to fully cover all the required bands, especially at the lower frequencies due to the narrower dual bandwidth, [3][4][5]9,16 or their size or thickness makes them difficult to integrate within mobile devices or portable wireless modules. 7,12,13,15 In this article, we propose novel multi-standard asymmetric SIR filters including both dual-wideband filters with wide stopbands and a quad-wideband filter. These filters are capable of generating two and four wide operating bands that effectively cover the GPS/GSM/UMTS/IEEE802.11a application in mobile devices, including GPS (1.75 GHz), GSM1800 (1710-1880 MHz), GSM1900 (1850-1990 MHz), UMTS (1920-2170 MHz), and IEEE 802.11a (5 GHz) bands. Meanwhile, in-band and out-of-band performance of the proposed filters is further enhanced by novel structures. To the best of authors' knowledge, these structures realize for the first time dual wideband and wide stop band resulting from the restriction of high order harmonic frequencies at the same time in dual wideband 8,10,11,[13][14][15] and asymmetric SIR filters. [3][4][5][6][7] Moreover, the proposed structures use the capacitive coupling of only two miniaturized resonators to realize dual-wideband and quad-wideband responses without such extra structures such as via holes or defected ground planes, and so are unique among dual-wideband and quadwideband filters. [10][11][12][13]15 The filters are simulated and optimized using CST microwave studio software. 23 2 | RESONANCE CHARACTERISTICS OF THE ASYMMETRIC SIR UNIT AND THE SKEW-SYMMETRICAL ASYMMETRIC SIR COUPLED PAIR

| Characteristic of the asymmetric SIR unit
The asymmetric SIR shown in Figure 1, adopted from Reference 4, consists of sections with low and high characteristic impedances Z 1 and Z 2 . The physical lengths L 1 and L 2 , physical widths W 1 and W 2 , and electrical lengths θ 1 and θ 2 are shown for the two sections with Z 1 and Z 2 , respectively. The characteristic impedance ratio K and the length ratio α are defined as follows: where α is located in the range of (0, 1). The input admittance Y in of the proposed asymmetric SIR unit is derived as: It is known that resonance of the proposed asymmetric SIR occurs when Y in = 0. Based on Equation (3), this resonance happens when From the solution of Equation (4), the first and second spurious frequencies f s1 and f s2 are plotted in Figure 2, normalized by the fundamental frequency against α with different values of K. When α is more than 0.5, the normalized F I G U R E 1 Structure of an asymmetric SIR 4 frequency f s1 /f 0 is greater than 2 and f s2 /f 0 is greater than 3, respectively. Also smaller K can result in greater normalized frequency when α is fixed. When α is greater than 0.5, the normalized frequency f s1 /f 0 is less than 2 and f s2 /f 0 is less than 3. Smaller K can result in lower normalized frequency for a fixed α.When K = 1, f s1 /f 0 is equal to 2 and f s2 /f 0 is equal to 3. This means that in this limit a uniform impedance resonator is realized and the high order resonant frequency is an integer multiple of the fundamental frequency f 0 . Therefore, the higher order spurious resonant modes, which depend on the choice of the characteristic impedance ratio K and the electric length ratio α, can be found by combining Equations (2) and (4).

| Characteristic of the skew-symmetrical asymmetric SIR coupled pair
A skew-symmetrical asymmetric SIR coupled pair is illustrated in Figure 3. It contains two asymmetric SIR units connected through two high impedance coupled lines.
The coupling matrix referred to in Reference 16 will not be discussed in this article because of its non-wideband limitation. 16,17 The coupling between two ASIRs can be represented by a J-inverter susceptance J 1,2 where the subscript 1 and 2 denotes the first and second passband. A larger value of J 1, 2 means a stronger coupling strength between two ASIRs. The normalized J 1,2 can be determined by where Z 0 represents the referred port impedance. The external quality factor Q ex1,2 and the normalized J-inverter susceptance J 1, 2 can be related by Reference 16.
From the full-wave EM simulation (CST software 23 ), it is noted that the frequency response performance (in terms of F I G U R E 2 f s1 (the first spurious frequency) and f s2 (the second spurious frequency) normalized by f 0 (the fundamental frequency) for the asymmetric SIR in Figure 1 F I G U R E 3 Structure of a skew-symmetrical asymmetric SIR coupled pair bandwidth, return loss and insertion loss) is improved significantly at some frequency points in the SS-ASIR coupled pair while in other respects the performance is not changed much or degraded in comparison with the asymmetric SIR unit. Table 1 shows the frequency response transformation from asymmetric SIR unit to skew-symmetrical asymmetric SIR coupled pair. It can be seen that when the asymmetric SIR's electric length ratio α ranges from 0.4 to 0.5, the frequency response at frequency f 0 , f s3 and f s4 enhances significantly in the SS-ASIR coupled structure, while it does not change much or degrades at f s1 and f s3 . When α = 0.6, the frequency response at f 0 , f s1 , and f s3 enhances greatly, but it does not vary much or degrades at f s2 and f s4 . The transformation table is useful to analysis and transform frequency bands where the performance-enhanced frequency response appears into desired pass bands by considering the asymmetric SIR unit characteristic. It is also useful to analyze and transform frequency bands where the performance-degraded frequency response appears into stop bands or excite them to become pass band by modifying the SS-ASIR coupled structure.

| The influence of electric length ratio α in SS-ASIRs
An inherent property of the skew-symmetrical asymmetric SIR (SS-ASIR) coupled pair is that the resonant frequency location and bandwidth performance are mainly influenced by the electric length ratio α when length L 1 is fixed, and is hardly affected by the impedance ratio K. This invariance characteristic is not found with the asymmetric SIR unit, where the normalized frequency (and resonant frequency) is closely related with K value. This characteristic of α = 0.42 is shown in Figure 4: resonant frequency locations and their bandwidths are nearly fixed when α is fixed at 0.42, and these two parameters are little influenced by K. Similar phenomenon can be observed when α is fixed at other values. This means α is the main factor to influence the horizontal frequency response performance such as resonant frequency locations and bandwidths in SS-ASIR coupled structures.

| The influence of impedance ratio K in SS-ASIRs
The return loss and insertion loss performance of enhanced f 0 and f s2 (seen in Table 1 and Figure 4) becomes better when K varies from 0.3 to 0.6. Meanwhile, the return loss and insertion loss performance of degraded f s1 and f s3 (seen in Table 1 and Figure 4) become better and forms two spurious peaking finally. Similar phenomenon can be observed when K takes other values. Therefore, the impedance ratio K is a major influence on the vertical frequency response performance such as return loss and insertion loss in both pass-band and stop-band of SS-ASIR coupled structures. By considering the characteristic of SS-ASIR coupled pairs and of the asymmetric SIR unit, α = 0.42 and K = 0.48 are extracted to design the proposed filters. The performance-enhanced frequency bands at f 0 and f s2 are used to form the first and second pass band in the proposed dual-band and quad-band filters. The performance-degraded frequency band at f s3 and performance-enhanced frequency band at f s4 are unwanted in the proposed dual-band filters, but they are used to form the third and fourth pass band in the proposed quad-band filter. Figure 5 plots the variation of coupling bandwidth as a function of the gap size (S 1 ) between the two resonators. Increasing the gap between the resonators reduces the coupling bandwidth of the designed filer and vice versa. Given the required electrical length ratio α, coupling coefficients and external quality factor for the proposed filters, one may determine the proper specifications based on these factors.
The transformation relationship of in-band performance of skew-symmetrical asymmetric SIR coupled pair and asymmetric SIR unit. "Improved" means significant enhancement of performance at relative frequency; "NC/DE" means no significant change or degradation of performance at relative frequency point. When α ranges from 0. 3

| SS-ASIR filter with open stub and folded coupled lines
In the skew-symmetrical asymmetric SIR filter, open stubs can be added to the low impedance lines, and the high impedance coupled lines can be folded, to achieve optimized performance. This geometry and its equivalent circuit are shown in Figures 6 and 7, respectively. When two open stubs are moved from point A to C, the suppression performance of f s3 and f s4 becomes progressively better while the return loss performance of f s2 become worse from A to B and better from B to C. Meanwhile, the insertion loss and return loss performance of fundamental frequency f o remains almost the same. Therefore, two open stubs are placed at point C. Figure 8 shows the open stub effect on the frequency response of the SS-ASIR filter.
The high impedance lines are also folded to form the section of length H 1 between the high and low impedance lines. Compared to conventional coupled lines that are not folded, the suppression of unwanted transmission at frequency f s4 is greatly improved when the interval H 1 varies from 0.5 mm to 0.9 mm. The change in the frequency response for this variation of H 1 is plotted in Figure 9.
The design procedures for dual-and quint-wideband type BPFs can be summarized as follows: 1. Choose a suitable electrical length ratio α, thus setting the fundamental frequency f 0 , and choose the characteristic impedance ratio K in the ASIR to improve insertion loss and return loss performance. 2. Analyze the transmission zero generating requirement of the interdigital cross-coupled line section added to the SS-ASIR structure and find suitable transmission zero locations. 3. According to previous results, tune the length of the interdigital cross-coupled line section to meet the required S 21 to form a wide stop band for the dualwideband type ASIR filter. The gap parameter S 5 is also tuned for optimized results.

4.
Tune the length of the interdigital cross-coupled line section to enable a multi-band response with good isolation between operating bands.

5.
The same design steps can be applied for the quintwideband type BPF.
Because of the non-wideband limitation of the coupling matrix, coupling coefficients are not important in this design, while the external quality factor Q ex can be discussed for performance optimization, as mentioned above.

| The modified SS-ASIR filter with interdigital cross-coupled line (ICCL)
To further optimize the in-band and out-of-band performance of the proposed dual-band filter, auxiliary interdigital cross-coupled lines whose width and length are W 5 and L 5 are utilized to realize multi-path coupling between two  Figure 10. Compared to the single coupling route of A-C, at least two extra coupling routes including A-D and B-C are created by the inclusion of the auxiliary coupled lines. This interdigital multipath coupling scheme of the modified SS-ASIR filter is shown in Figure 10B. Moreover, because of the small distance between the main and auxiliary coupled lines, mutual coupling of A-B and C-D exists at the same time. This coupling is illustrated as dashed gray lines in Figure 10B.
By including auxiliary coupled lines, the filter's out-ofband spurious frequency suppression performance and passband selectivity is considerably improved. The designed dual-band band pass filter (BPF) adopting an interdigital cross-coupled configuration produces a transmission zero (TZ) to approach f s3 and f s4 . The transmission zero occurs because of cancelation of the transmitted signals passing through different routes. The transmission line equivalent circuit model and the coupling routing scheme for the SS-ASIRs are shown in Figure 10C. The impedance values of the transmission lines are: Z 1 = 73.4856 Ω, Z 2 = 23.5845 Ω, Z A = Z C = 67.0065 Ω, and Z B = Z D = 73.5209 Ω, these values have been achieved and verified with the aid of the spice extraction tool embedded in the CST software. 17,23 As seen in Figure 11, when the length of auxiliary coupled line L 5 ranges from 1.4 to 1.8 mm, optimized suppression of f s3 and f s4 is achieved. Also a wide stop band ranging from 6.3 to 12 GHz at the upper side of the second pass band is realized simultaneously. Also, the second pass band upper side's selectivity is improved with the two pass bands' return loss and insertion loss performance unaffected. Figure 12 plots Q ex1 , Q ex2 , f s2 /f 0 and Q ex2 /Q ex1 vs S 5 , which is the gap between auxiliary coupled line and main coupled line in the interdigital cross-coupled SS-ASIR filter. When  Figure 4. Q ex2 decreases initially and increases a little later: this means that the bandwidth of f s2 can be controlled and expanded by tuning S 5 . Therefore, S5 is a factor that can tune Q ex2 and J 1, 2 in SS-ASIRs with the interdigital cross coupled line structure.
The proposed filters were fabricated on an RO3010 substrate with a relative permittivity of 10.2, and measured using an HP8550 vector network analyzer. The simulated S-parameters, measured S-parameters and photograph of the hardware realization of the designed dual-wideband SS-ASIRs with ICCLs are plotted in Figure 13. Good agreement is observed between the simulated and measured results and the slight discrepancies are attributed to losses and fabrication errors. Dual wide bands are realized with good in-band return loss performance. The first pass band ranges from 1.  1 GHz with −10 dB suppression is generated, which can be seen in Figure 13. Parameters of the modified SS-ASIR filter and modified SS-ASIR filter with interdigital cross coupled lines (ICCLs) are shown in Table 2.

| The modified SS-ASIR filter with parallel uncoupled microstrip lines (PUMLs)
An uncoupled section located within conventional coupled lines is a useful method to achieve extra transmission zeros close to existing zeros created by the conventional coupled lines. Therefore, the results extending the stopband bandwidth with a better suppression level can be achieved. At the same time, this method can also give freedom to optimize the in-band performance of the original structure. Figure 14 shows the topological structure of the proposed modified SS-ASIR filter with parallel uncoupled microstrip lines. These parallel uncoupled microstrip lines are formed by bending the original coupled lines outwards. The parallel uncoupled microstrip line's reference location to the original coupled line open end is L r . The parallel uncoupled microstrip line height and inner gap is L m and W m , respectively. The same procedure applied to achieve the equivalent circuit  Figure 10C, is adopted for the SS-ASIR filter with PUMLs as shown in Figure 14B. The impedance values of the transmission lines are: Z 1 = 73.4856 Ω, Z 2 = 23.5845 Ω, Z 3 = 67.4162 Ω, and Z 4 = 58.6799 Ω.
Compared to the former modified SS-ASIR filter with its interdigital cross-coupled line structure, the wider second pass band is achieved by adopting novel parallel uncoupled microstrip lines. In the former structure, Q ex1 and Q ex2 are equal to 2.28 and 9.37 when S 5 = 0.25 mm (as shown in Figure 11), while Q ex1,2 equal 2.71 and 3.48 when L m = 2.3 mm in this novel filter with PUMLs. This means J 2 is improved greatly and a stronger coupling strength between two modified ASIRs is realized. Figure 15 shows the impact of the reference location parameter L r on the frequency response of the filter with parallel uncoupled microstrip lines. It is noted that when L r changes from 5.6 mm to 7 mm, the second pass band return loss performance is enhanced considerably and its bandwidth becomes wider resulting in forming a wide second pass band of more than 1.5 GHz. Meanwhile, the frequency response of the first pass band does not vary much. Hence, L r is a vital factor to influence J 2 and Q ex2 . Moreover, the parallel uncoupled microstrip line structure achieves one extra transmission zero nearby f s3 and f s4 , leading to an extended wide stopband bandwidth with better suppression level, as seen in Figure 15. Figures 16 and 17 show the impact of L m on the frequency response of the SS-ASIR filter with PUMLs. In Figure 16, when L m ranges from 0.5 mm to 5 mm, the fundamental frequency f 0 decreases continuously while Q ex1 increases slightly at first and increases dramatically when L m is greater than 2.5 mm. This shows that the parallel uncoupled microstrip line height L m can increase Q ex1 and decrease J 2 within a certain range. Figure 17 plots the f s2 , Q ex2 , f s2 /f 0 and Q ex2 /Q ex1 vs against L m . When L m ranges from 0.5 mm to 1.5 mm, the rate of decline of Q ex2 is much faster than the rate of decline of f s2 , which means the bandwidth centered at f s2 is growing rapidly. When L m ranges from 1.5 mm to 2.5 mm, the rate of decline of Q ex2 is almost the same as that of f s2 , that means that a wide band width centered at f s2 is formed and does not change much. When L m ranges from 2.5 mm to 5 mm, Q ex2 increases as f s2 decreases, which means that the band centered at f s2 becomes narrower. In the whole process, f s2 moves from F I G U R E 1 7 f s2 , Q ex2 , f s2 / f 0 and Q ex2 /Q ex1 vs against L m F I G U R E 1 8 The analysis of the PUML unit 5.6 GHz to 3.8 GHz with the second pass band being expanded to more than 1.5 GHz. Hence, L m is also a factor to influence Q ex2 and strengthen J 2 .
The reason for this can be explained by analyzing the parallel uncoupled microstrip line unit, whose topological structure and frequency response are plotted in Figure 18. As seen in the figure, the PUML unit forms a wide pass band of more than 1 GHz when L m changes from 1 mm to 3 mm, and the pass band central frequency can be tuned by L m . This result proves the advantage of the PUML structure to optimize the in-band performance of the filter. As for outof-band performance, the PUML unit generates three transmission zeros (TZs) at both sides of the pass band, as plotted in Figure 18. These three TZs can improve the isolation performance between two pass bands and the suppression level of undesired f s3 and f s4 .
The simulated S-parameters, measured S-parameters and photograph of the hardware realization of the designed dualwideband SS-ASIRs with PUMLs are plotted in Figure 19. There is good agreement between the simulated and measured results and the slight discrepancies are attributed to losses and fabrication errors. It can be seen that dual wide bands are realized with good in-band return loss performance. The first pass band ranges from 1.37 to 1.  5.83 to 9.35 GHz with −10 dB suppression level is generated, which can be seen in Figure 19. The parameters of the modified SS-ASIR with parallel uncoupled microstrip lines are shown in Table 2. A comparison of the performance of the modified SS-ASIR filter with ICCLs and PUMLs with previous work is shown in Table 3.

| The modified SS-ASIR filter with defected rectangular structure (DRS)
The topological structure of the proposed modified skewsymmetrical asymmetric SIR Filter with defected structure is  Figure 20. In contrast to the original skewsymmetrical asymmetric SIR filter, this filter has a defected rectangular structure (DRS) in each low impedance line, of width and length W d and H d , respectively. Its relative distance to the end of the feed line is R d . By utilizing the defected rectangular structure, the performance-degraded frequency response at f s3 can enhance the frequency response and a new wide pass band can be excited centered at f s3 . The frequency response comparison between the modified SS-ASIR filter with and without DRS is plotted in Figure 20C. By this means, a quad-band filter can be formed by adopting the defected rectangular structure in the SS-ASIR filter.
Again, the same procedure applied to achieve the equivalent circuit elements used in Figures 10C and 14B, is adopted for the SS-ASIR filter with DRS and without DRS as shown in Figure 20B. The impedance values of the transmission lines are: Z 1 = 34.6069 Ω, Z 2 = 49.6435 Ω, Z 3 = 23.5845 Ω, and Z 4 = 67.0065 Ω. Figure 21 plots the W d 's effect on SS-ASIR filter with DRS. In (a), when W d varies from 2.5 to 6 mm, f s2 /f 0 decreases while Q ex1 remains almost the same: Q ex2 and Q ex2 /Q ex1 fluctuate in this process. This means f s2 does not follow the rules of invariance of resonant frequency location in the SS-ASIR coupled structure any more. In (b), Q ex3 varies from 12.1 to 26.8 and Q ex4 varies from 18.1 to 16 when W d ranges from 2.5 to 6 mm, with Q ex3 /Q ex1 increasing continuously.
From EM simulations, it is noted that not all f s2 , f s3 , and f s4 in SS-ASIR coupled structures with DRS obey the rules of invariance of resonant frequency location: frequency shifting exists among all four pass bands compared to the situation that there is no DRS. This is because the basic structure of SS-ASIR coupled pairs is modified by the defected rectangular structure.
The simulated S-parameters, measured S-parameters and photograph of the hardware realization of the designed quad-wideband SS-ASIR filter with DMS are plotted in Figure 22. There is good agreement between the simulated and measured results and the discrepancies are attributed to loss, fabrication errors and so on. It can be seen that dual wide bands are realized with good in-band return loss performance. The first pass band ranges from 1. Moreover, good isolation is achieved between four pass bands, eliminating signal interference. The −10 dB suppression level stop bands range from 3.05 to 4.78 GHz between the first and second pass band, 5.77 to 7.20 GHz between the second and third pass band, and 8.04 to 8.889 GHz between the third and fourth pass band. Moreover, two transmission zeroes located at 4.26 GHz and 8.46 GHz are formed to further enhance frequency selectivity, as shown in F I G U R E 2 1 W d 's effect on SS-ASIR filter with DRS. (A) Q ex1 , Q ex2 , fs2/f 0 and Q ex2 /Q ex1 vs against W d . (B) Q ex3 , Q ex4 , Q ex3 /Q ex1 and Q ex4 /Q ex1 vs against W d Figure 22. The parameters of modified SS-ASIR with DMS are shown in Table 2. The size of the modified SS-ASIRs with DMS is 0.09λ g × 0.56λ g , where λ g is the guide wavelength.

| Surface current distribution for the proposed filters
For further validation by using the CST software simulator, Figure 23 elaborates the current distribution of the reported pass band filters at the resonant frequencies.
As shown in Figure 23A, the electric field is mainly distributed near the interdigital cross-coupled lines, with maximum current density 60 and 58 A/m at 1.8 and 5.5 GHz, respectively. But for Figure 23B, the electric field is mainly distributed near the parallel uncoupled microstrip lines, with maximum current densities 58 and 57 A/m at 1.6 and 4.4 GHz, respectively. On the other hand, as seen from Figure 23C, most of the currents are around the defected rectangular structures and the two coupled transmission lines with maximum current densities 60 A/m at 2.1, 5.2, 7.6, and 9.3 GHz. It is significant that the proposed designs can be easily developed to handle and permit reconfigurability 24,25 and can be easily integrated with antenna designs, 26 to create the so-called "filtenna". 27

| CONCLUSION
Multi-standard dual-wideband and quad-wideband filters based on the detailed analysis of the simple asymmetric stepped-impedance resonator unit and of the skewsymmetrical asymmetric stepped-impedance resonator coupled pair have been presented. By utilizing a novel modified skew-symmetrical asymmetric stepped-impedance resonator coupled pair with interdigital cross-coupled lines and parallel uncoupled microstrip lines, the dual-band filters for the first time among recently proposed dual-wideband filters realize dual-wideband with wide stop band restrictions. Their bandwidths are controllable by tuning relative parameters. By introducing a defected microstrip structure, a frequency band nearby the third spurious frequency is formed, resulting in a quad-wideband filter. The proposed dualwideband and quad-wideband modified skew-symmetrical asymmetric stepped-impedance resonator filters cover communication applications including Global Positioning System, Global System For Mobile Communication, The Universal Mobile Telecommunications System, industrial, scientific and medical band, and IEEE 802.11 a/b/g/n/ac. The filters' measured results agree well with simulated results and theoretical predictions. The good in-band and out-of-band performance, compact size and simple structure make the proposed filters very promising for applications in future multi-standard wireless communication.

ACKNOWLEDGMENTS
This project has received funding from the European Union's Horizon 2020 research and innovation programme under grant agreement H2020-MSCA-ITN-2016 SECRET-722424. worked as Lecturer at University of Hail, Saudi Arabia. Now he is currently working toward his PhD degree within the radio frequency and sensor design research group. His research interests include simulation, design, and implementation of front-end antenna systems for millimeter wave communications.

ORCID
Raed A. Abd-Alhameed (M'02-SM'13) received the BSc and MSc degrees in Electrical Engineering from Basrah University, Basrah, Iraq, in 1982 and 1985, respectively, and the PhD degree in Electrical Engineering from the University of Bradford, UK, in 1997. He has also been a Research Visitor with Glyndwr University, Wrexham, UK, since 2009. He is currently a Professor of electromagnetic and radio frequency engineering with the University of Bradford. He is the Leader of radio frequency, propagation, sensor design, and signal processing. He leads the Communications Research Group, School of Engineering and Informatics, Bradford University, for years. He is investigator of several funded applications. He has published more than 500 academic journal and conference papers, and books. His research interests include computational methods and optimizations, wireless and mobile communications, sensor design, signal processing, EMC, propagations, antennas, beam steering, energy-efficient PAs, and RF predistorter design applications.
James M. Noras is currently a Senior Lecturer with the School of Engineering, Design and Technology, University of Bradford, UK. He is also the Director of five internationally franchised BEng and MSc courses in electrical and electronic engineering, has successfully supervised 18 PhD students, and is currently supervising the research of 3 PhD students. He has published 50 journal papers and 85 conference papers, in fundamental semiconductor physics, analog and digital circuit design, digital signal processing, and RF system design and evaluation. His main research interests include digital system design and implementation, DSP and coding for communication systems, and localization algorithms for mobile systems. He is a member of the Institute of Physics and a Chartered Physicist.